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1 ltc1625 no r sense tm current mode synchronous step-down switching regulator features descriptio n u n highest efficiency current mode controller n no sense resistor required n stable high current operation n dual n-channel mosfet synchronous drive n wide v in range: 3.7v to 36v n wide v out range: 1.19v to v in n 1% 1.19v reference n programmable fixed frequency with injection lock n very low drop out operation: 99% duty cycle n forced continuous mode control pin n optional programmable soft start n pin selectable output voltage n foldback current limit n output overvoltage protection n logic controlled micropower shutdown: i q < 30 m a n available in 16-lead narrow ssop and so packages the ltc ? 1625 is a synchronous step-down switching regulator controller that drives external n-channel power mosfets using few external components. current mode control with mosfet v ds sensing eliminates the need for a sense resistor and improves efficiency. the frequency of a nominal 150khz internal oscillator can be synchronized to an external clock over a 1.5:1 frequency range. burst mode tm operation at low load currents reduces switching losses and low dropout operation extends oper- ating time in battery-powered systems. a forced continu- ous mode control pin can assist secondary winding regulation by disabling burst mode operation when the main output is lightly loaded. fault protection is provided by foldback current limiting and an output overvoltage comparator. an external ca- pacitor attached to the run/ss pin provides soft start capability for supply sequencing. a wide supply range allows operation from 3.7v (3.9v for ltc1625i) to 36v at the input and 1.19v to v in at the output. typical applicatio n u figure 1. high efficiency step-down converter + + v in tk sync ltc1625 run/ss v osense tg sw c b 0.22 m f d b cmdsh-3 c c 2.2nf r c 10k m2 si4410dy d1 mbrs140t3 m1 si4410dy c vcc 4.7 m f 1625 f01 l1 10 m h c in 10 m f 30v 2 boost intv cc bg i th v prog sgnd pgnd + c out 100 m f 10v 3 v out 3.3v 4.5a v in 5v to 28v c ss 0.1 m f load current (a) 0.01 efficiency (%) 80 90 100 0.1 1 10 1625 ta01 70 60 v in = 10v v out = 5v v out = 3.3v efficiency vs load current applicatio n s u n notebook and palmtop computers, pdas n cellular telephones and wireless modems n battery chargers n distributed power , ltc and lt are registered trademarks of linear technology corporation. no r sense and burst mode are trademarks of linear technology corporation.
2 ltc1625 absolute m axi m u m ratings w ww u (note 1) input supply voltage (v in , tk) ................. 36v to C 0.3v boosted supply voltage (boost) ............. 42v to C 0.3v boosted driver voltage (boost C sw) ...... 7v to C 0.3v switch voltage (sw).....................................36v to C 5v extv cc voltage ........................................... 7v to C 0.3v i th voltage ................................................2.7v to C 0.3v fcb, run/ss, sync voltages .....................7v to C 0.3v v osense , v prog voltages ........(intv cc + 0.3v) to C 0.3v peak driver output current < 10 m s (tg, bg) ............ 2a intv cc output current ........................................ 50ma operating ambient temperature range LTC1625C............................................... 0 c to 70 c ltc1625i (note 5) .............................. C 40 c to 85 c junction temperature (note 2) ............................. 125 c storage temperature range ................ C 65 c to 150 c lead temperature (soldering, 10 sec)................. 300 c wu u package / o rder i for atio t a = 25 c, v in = 15v unless otherwise noted. electrical characteristics symbol parameter conditions min typ max units main control loop i in v osense feedback current v prog pin open, i th = 1.19v (note 3) 10 50 na v out regulated output voltage i th = 1.19v (note 3) 1.19v (adjustable) selected v prog pin open l 1.178 1.190 1.202 v 3.3v selected v prog = 0v l 3.220 3.300 3.380 v 5v selected v prog = intv cc l 4.900 5.000 5.100 v v linereg reference voltage line regulation v in = 3.6v to 20v, i th = 1.19v (note 3), 0.001 0.01 %/v v prog pin open v loadreg output voltage load regulation i th = 2v (note 3) l C 0.020 C 0.2 % i th = 0.5v (note 3) l 0.035 0.2 % v fcb forced continuous threshold v fcb ramping negative l 1.16 1.19 1.22 v i fcb forced continuous current v fcb = 1.19v C 1 C 2 m a v ovl output overvoltage lockout v prog pin open 1.24 1.28 1.32 v i prog v prog input current 3.3v v out v prog = 0v C 3.5 C 7 m a 5v v out v prog = 5v 3.5 7 m a i q input dc supply current extv cc = 5v (note 4) normal mode 500 m a shutdown v run/ss = 0v, 3.7v < v in < 15v 15 30 m a v run/ss run/ss pin threshold l 0.8 1.4 2 v i run/ss soft start current source v run/ss = 0v 1.2 2.5 4 m a d v sense(max) maximum current sense threshold v osense = 1v, v prog pin open 120 150 170 mv tg transition time tg t r rise time c load = 3300pf 50 150 ns tg t f fall time c load = 3300pf 50 150 ns order part number LTC1625Cgn LTC1625Cs ltc1625ign ltc1625is consult factory for military grade parts. top view s package 16-lead plastic so gn package 16-lead plastic ssop 1 2 3 4 5 6 7 8 16 15 14 13 12 11 10 9 extv cc sync run/ss fcb i th sgnd v osense v prog v in tk sw tg boost intv cc bg pgnd t jmax = 125 c, q ja = 130 c/w (gn) t jmax = 125 c, q ja = 110 c/w (s)
3 ltc1625 t a = 25 c, v in = 15v unless otherwise noted. electrical characteristics symbol parameter conditions min typ max units bg transition time bg t r rise time c load = 3300pf 50 150 ns bg t f fall time c load = 3300pf 50 150 ns internal v cc regulator v intvcc internal v cc voltage 6v < v in < 30v, v extvcc = 4v l 5.0 5.2 5.4 v v ldoint intv cc load regulation i cc = 20ma, v extvcc = 4v C 1 C 2 % v ldoext extv cc voltage drop i cc = 20ma, v extvcc = 5v 180 300 mv v extvcc extv cc switchover voltage i cc = 20ma, v extvcc ramping positive l 4.5 4.7 v oscillator f osc oscillator freqency 135 150 165 khz f h /f osc maximum synchronized frequency ratio 1.5 v sync sync pin threshold (figure 4) ramping positive 0.9 1.2 v r sync sync pin input resistance 50 k w the l denotes specifications which apply over the full operating temperature range. note 1: absolute maximum ratings are those values beyond which the life of a device may be impaired. note 2: t j is calculated from the ambient temperature t a and power dissipation p d according to the following formula: LTC1625Cgn/ltc1625ign: t j = t a + (p d ? 130 c/w) LTC1625Cs/ltc1625is: t j = t a + (p d ? 110 c/w) note 3: the ltc1625 is tested in a feedback loop that adjusts v osense to achieve a specified error amplifier output voltage (i th ). note 4: typical in application circuit with extv cc tied to v out = 5v, i out = 0a and fcb = intv cc . dynamic supply current is higher due to the gate charge being delivered at the switching frequency. see applications information. note 5: minimum input supply voltage is 3.9v at C 40 c for industrial grade parts.
4 ltc1625 typical perfor a ce characteristics uw input voltage (v) 0 70 efficiency (%) 75 80 85 90 100 5 10 15 20 1625 g02 25 30 95 i load = 2a i load = 200ma figure 1 circuit efficiency vs input voltage, v out = 3.3v v in C v out dropout voltage vs load current load current (a) 0 v in ?v out (mv) 200 300 4 1625 g06 100 0 1 2 3 5 400 figure 1 circuit v out = 5v ?5% drop efficiency vs load current load current (a) 0.001 efficiency (%) 70 80 10 1625 g01 60 50 0.01 0.1 1 100 90 burst mode operation continuous mode v in = 10v v out = 5v extv cc = v out load regulation load current (a) 0 d v out (%) 0.10 0.05 0 4 1625 g04 0.15 0.20 0.25 1 2 3 5 figure 1 circuit input and shutdown current vs input voltage input voltage (v) 05 0 input current ( m a) shutdown current ( m a) 400 1000 10 20 25 1625 g07 200 800 600 0 20 50 10 40 30 15 30 35 extv cc open extv cc = 5v shutdown efficiency vs input voltage, v out = 5v input voltage (v) 0 70 efficiency (%) 75 80 85 90 100 5 10 15 20 1625 g02 25 30 95 i load = 2a i load = 200ma figure 1 circuit i th pin voltage vs load current load current (a) 0 v ith (v) 2.0 2.5 3.0 35 1625 g05 1.5 1.0 12 467 0.5 0 figure 1 circuit v in = 20v v out = 5v continuous mode burst mode operation intv cc load current (ma) 0 extv cc ?intv cc (mv) 300 400 500 40 1625 g09 200 100 0 10 20 30 50 extv cc switch drop vs intv cc load current intv cc load regulation intv cc load current (ma) 0 d intv cc (%) ?.0 0.5 0 40 1625 g08 ?.5 2.0 2.5 10 20 30 50
5 ltc1625 typical perfor a ce characteristics uw oscillator frequency vs temperature fcb pin current vs temperature maximum current sense voltage vs temperature maximum current sense voltage vs duty cycle duty cycle 0 maximum current sense voltage (mv) 100 150 0.8 1625 g10 50 0 0.2 0.4 0.5 1.0 200 temperature ( c) ?0 140 maximum current sense voltage (mv) 145 150 155 160 ?5 10 35 60 1625 g11 85 110 135 temperature ( c) ?0 frequency (khz) 200 250 300 35 85 1625 g12 150 100 ?5 10 60 110 135 50 0 sync = 1.5v sync = 0v temperature ( c) ?0 fcb current ( m a) 0.50 0.25 0 35 85 1625 g13 0.75 1.00 ?5 10 60 110 135 1.25 1.50 run/ss pin current vs temperature temperature ( c) ?0 ?5 ? run/ss current ( m a) ? 0 10 60 85 1625 g14 ? ? ? 35 110 135 soft start: load current vs time inductor current 2a/div run/ss 2v/div 20ms/div v in = 20v v out = 5v r load = 1 w figure 1 circuit v in = 20v v out = 5v i load = 1a to 4a figure 1 circuit v out 50mv/div 200 m s/div 50 m s/div v in = 20v v out = 5v i load = 50ma figure 1 circuit burst mode operation v out 50mv/div i th 100mv/div transient response (burst mode operation) v out 50mv/div 500 m s/div v in = 20v v out = 5v i load = 50ma to 1a figure 1 circuit transient response 1625 f07 1625 f09 1625 f06 1625 f08
6 ltc1625 pi n fu n ctio n s uuu leaving v prog open allows the output voltage to be set by an external resistive divider between the output and v osense . pgnd (pin 9): driver power ground. connects to the source of the bottom n-channel mosfet, the (C) terminal of c vcc and the (C) terminal of c in . bg (pin 10): bottom gate drive. drives the gate of the bottom n-channel mosfet between ground and intv cc . intv cc (pin 11): internal 5.2v regulator output. the driver and control circuits are powered from this voltage. decouple this pin to power ground with a minimum of 4.7 m f tantalum capacitance. boost (pin 12): topside floating driver supply. the (+) terminal of the bootstrap capacitor connects here. this pin swings from a diode drop below intv cc to v in + intv cc . tg (pin 13): top gate drive. drives the top n-channel mosfet with a voltage swing equal to intv cc minus a diode drop, superimposed on the switch node voltage. sw (pin 14): switch node. the (C) terminal of the boot- strap capacitor connects here. this pin swings from a diode drop below ground up to v in . tk (pin 15): top mosfet kelvin sense. mosfet v ds sensing requires this pin to be routed to the drain of the top mosfet separately from v in . v in (pin 16): main supply input. decouple this pin to ground with an rc filter (4.7 w , 0.1 m f) for applications above 3a. extv cc (pin 1): intv cc switch input. when the extv cc voltage is above 4.7v, the switch closes and supplies intv cc power from extv cc . do not exceed 7v at this pin. sync (pin 2): synchronization input for internal oscilla- tor. the oscillator will nominally run at 150khz when open, 225khz when tied above 1.2v, and will lock over a 1.5:1 clock frequency range. run/ss (pin 3): run control and soft start input. a capacitor to ground at this pin sets the ramp time to full current output (approximately 1s/ m f). forcing this pin below 1.4v shuts down the device. fcb (pin 4): forced continuous input. tie this pin to ground to force synchronous operation at low load, to a resistive divider from the secondary output when using a secondary winding, or to intv cc to enable burst mode operation at low load. i th (pin 5): error amplifier compensation point. the current comparator threshold increases with this control voltage. nominal voltage range for this pin is 0v to 2.4v. sgnd (pin 6): signal ground. connect to the (C) terminal of c out . v osense (pin 7): output voltage sense. feedback input from the remotely sensed output voltage or from an external resistive divider across the output. v prog (pin 8): output voltage programming. when v osense is connected to the output, v prog < 0.8v selects a 3.3v output and v prog > 3.5v selects a 5v output.
7 ltc1625 fu n ctio n al diagra uu w + intv cc c vcc + c in m2 v in 16 extv cc fcb 1 4 v prog 8 sgnd 6 run/ss 3 bg 10 pgnd 4.7v 1.19v c ss 1 m a l1 1625 bd 9 boost v in c b m1 d b 12 tk rev 15 sync 2 tg 13 sw 14 switch logic/ dropout counter + + + + ta 11 ba 11 i 2 1.19v ref fcnt overvoltage shutdown top 0.6v 5.2v ldo reg 1.28v 1.19v v fb + + + + + c out 3 m a 6v 0.6v g m = 1m w + cl v osense 7 + 0.6v 0.95v i th 5 c c1 r c sleep 0.5v i thb + + i 1 s osc q r b + f ov ea 11
8 ltc1625 operatio u main control loop the ltc1625 is a constant frequency, current mode controller for dc/dc step-down converters. in normal operation, the top mosfet is turned on when the rs latch is set by the on-chip oscillator and is turned off when the current comparator i 1 resets the latch. while the top mosfet is turned off, the bottom mosfet is turned on until either the inductor current reverses, as determined by the current reversal comparator i 2 , or the next cycle begins. inductor current is measured by sensing the v ds potential across the conducting mosfet. the output of the appropriate sense amplifier (ta or ba) is selected by the switch logic and applied to the current comparator. the voltage on the i th pin sets the comparator threshold corresponding to peak inductor current. the error ampli- fier ea adjusts this voltage by comparing the feedback signal v fb from the output voltage with the internal 1.19v reference. the v prog pin selects whether the feedback voltage is taken directly from the v osense pin or is derived from an on-chip resistive divider. when the load current increases, it causes a drop in the feedback voltage relative to the reference. the i th voltage then rises until the average inductor current again matches the load current. the internal oscillator can be synchronized to an external clock applied to the sync pin and can lock to a frequency between 100% and 150% of its nominal 150khz rate. when the sync pin is left open, it is pulled low internally and the oscillator runs at its normal rate. if this pin is taken above 1.2v, the oscillator will run at its maximum 225khz rate. pulling the run/ss pin low forces the controller into its shutdown state and turns off both mosfets. releasing the run/ss pin allows an internal 3 m a current source to charge up an external soft start capacitor c ss . when this voltage reaches 1.4v, the controller begins switching, but with the i th voltage clamped at approximately 0.8v. as c ss continues to charge, the clamp is raised until full range operation is restored. the top mosfet driver is powered from a floating boot- strap capacitor c b . this capacitor is normally recharged from intv cc through a diode d b when the top mosfet is turned off. as v in decreases towards v out , the converter will attempt to turn on the top mosfet continuously (dropout). a dropout counter detects this condition and forces the top mosfet to turn off for about 500ns every tenth cycle to recharge the bootstrap capacitor. an overvoltage comparator ov guards against transient overshoots and other conditions that may overvoltage the output. in this case, the top mosfet is turned off and the bottom mosfet is turned on until the overvoltage condi- tion is cleared. foldback current limiting for an output shorted to ground is provided by a transconductance amplifer cl. as v fb drops below 0.6v, the buffered i th input to the current comparator is gradually pulled down to a 0.95v clamp. this reduces peak inductor current to about one fifth of its maximum value. low current operation the ltc1625 is capable of burst mode operation at low load currents. if the error amplifier drives the i th voltage below 0.95v, the buffered i th input to the current com- parator will remain clamped at 0.95v. the inductor current peak is then held at approximately 30mv/r ds(on)(top) . if i th then drops below 0.5v, the burst mode comparator b will turn off both mosfets. the load current will be supplied solely by the output capacitor until i th rises above the 50mv hysteresis of the comparator and switch- ing is resumed. burst mode operation is disabled by comparator f when the fcb pin is brought below 1.19v. this forces continuous operation and can assist second- ary winding regulation. intv cc /extv cc power power for the top and bottom mosfet drivers and most of the internal circuitry of the ltc1625 is derived from the intv cc pin. when the extv cc pin is left open, an internal 5.2v low dropout regulator supplies the intv cc power from v in . if extv cc is raised above 4.7v, the internal regulator is turned off and an internal switch connects extv cc to intv cc . this allows a high efficiency source, such as the primary or a secondary output of the converter itself, to provide the intv cc power.
9 ltc1625 applicatio n s i n for m atio n wu u u the basic ltc1625 application circuit is shown in figure 1. external component selection is primarily determined by the maximum load current and begins with the selection of the sense resistance and power mosfets. because the ltc1625 uses mosfet v ds sensing, the sense resistance is the r ds(on) of the mosfets. the operating frequency and the inductor are chosen based largely on the desired amount of ripple current. finally, c in is selected for its ability to handle the large rms current into the converter and c out is chosen with low enough esr to meet the output voltage ripple specification. power mosfet selection the ltc1625 requires two external n-channel power mosfets, one for the top (main) switch and one for the bottom (synchronous) switch. important parameters for the power mosfets are the breakdown voltage v (br)dss , threshold voltage v gs(th) , on-resistance r ds(on) , reverse transfer capacitance c rss and maximum current i d(max) . the gate drive voltage is set by the 5.2v intv cc supply. consequently, logic level threshold mosfets must be used in ltc1625 applications. if low input voltage opera- tion is expected (v in < 5v), then sub-logic level threshold mosfets should be used. pay close attention to the v (br)dss specification for the mosfets as well; many of the logic level mosfets are limited to 30v or less. the mosfet on-resistance is chosen based on the required load current. the maximum average output cur- rent i o(max) is equal to the peak inductor current less half the peak-to-peak ripple current d i l . the peak inductor current is inherently limited in a current mode controller by the current threshold i th range. the corresponding maximum v ds sense voltage is about 150mv under nor- mal conditions. the ltc1625 will not allow peak inductor current to exceed 150mv/r ds(on)(top) . the following equation is a good guide for determining the required r ds(on)(max) at 25 c (manufacturers specification), al- lowing some margin for ripple current, current limit and variations in the ltc1625 and external component values: r mv i ds on max o max t ()( ) () @ ()() 120 r the r t is a normalized term accounting for the significant variation in r ds(on) with temperature, typically about 0.4%/ c as shown in figure 2. junction to case tempera- ture t jc is around 10 c in most applications. for a maximum ambient temperature of 70 c, using r 80 c @ 1.3 in the above equation is a reasonable choice. this equation is plotted in figure 3 to illustrate the dependence of maximum output current on r ds(on) . some popular mosfets from siliconix are shown as data points. junction temperature ( c) ?0 r t normalized on resistance 1.0 1.5 150 1625 f02 0.5 0 0 50 100 2.0 figure 2. r ds(on) vs temperature r ds(on) ( w ) 0 maximum output current (a) 6 8 10 0.08 1625 f03 4 2 0 0.02 0.04 0.06 0.10 si4420 si4410 si4412 si9936 figure 3. maximum output current vs r ds(on) at v gs = 4.5v the power dissipated by the top and bottom mosfets strongly depends upon their respective duty cycles and the load current. when the ltc1625 is operating in con- tinuous mode, the duty cycles for the mosfets are:
10 ltc1625 applicatio n s i n for m atio n wu u u top duty cycle v v bottom duty cycle vv v out in in out in = = the mosfet power dissipations at maximum output current are: p v v ir kv i c f p vv v ir top out in o max t top ds on in o max rss bot in out in o max t bot ds on = ? ? ? ? + = ? ? ? ? ()()() ()( )( )( )() ()()() () () () () () () () 2 2 2 r r both mosfets have i 2 r losses and the p top equation includes an additional term for transition losses, which are largest at high input voltages. the constant k = 1.7 can be used to estimate the amount of transition loss. the bottom mosfet losses are greatest at high input voltage or during a short circuit when the duty cycle is nearly 100%. operating frequency and synchronization the choice of operating frequency and inductor value is a trade-off between efficiency and component size. low frequency operation improves efficiency by reducing mosfet switching losses, both gate charge loss and transition loss. however, lower frequency operation requires more inductance for a given amount of ripple current. the internal oscillator runs at a nominal 150khz frequency when the sync pin is left open or connected to ground. pulling the sync pin above 1.2v will increase the fre- quency by 50%. the oscillator will injection lock to a clock signal applied to the sync pin with a frequency between 165khz and 200khz. the clock high level must exceed 1.2v for at least 1 m s and no longer than 4 m s as shown in figure 4. the top mosfet turn-on will synchronize with the rising edge of the clock. 0 1 m s4 m s 1625 f04 7v 1.2v figure 4. sync clock waveform inductor value selection given the desired input and output voltages, the inductor value and operating frequency directly determine the ripple current: d i v fl v v l out out in = ? ? ? ? ? ? ? ? ()( ) 1 lower ripple current reduces core losses in the inductor, esr losses in the output capacitors and output voltage ripple. thus, highest efficiency operation is obtained at low frequency with small ripple current. to achieve this, however, requires a large inductor. a reasonable starting point is to choose a ripple current that is about 40% of i o(max) . note that the largest ripple current occurs at the highest v in . to guarantee that ripple current does not exceed a specified maximum, the induc- tor should be chosen according to: l v fi v v out l max out in max 3 ? ? ? ? ? ? ? ? ()( ) () () d 1 burst mode operation considerations the choice of r ds(on) and inductor value also determines the load current at which the ltc1625 enters burst mode operation. when bursting, the controller clamps the peak inductor current to approximately: i mv r burst peak ds on () () = 30
11 ltc1625 applicatio n s i n for m atio n wu u u the corresponding average current depends on the amount of ripple current. lower inductor values (higher d i l ) will reduce the load current at which burst mode operation begins. the output voltage ripple can increase during burst mode operation if d i l is substantially less than i burst . this will primarily occur when the duty cycle is very close to unity (v in is close to v out ) or if very large value inductors are chosen. this is generally only a concern in applications with v out 3 5v. at high duty cycles, a skipped cycle causes the inductor current to quickly descend to zero. however, it takes multiple cycles to ramp the current back up to i burst(peak) . during this interval, the output capaci- tor must supply the load current and enough charge may be lost to cause significant droop in the output voltage. it is a good idea to keep d i l comparable to i burst(peak) . otherwise, one might need to increase the output capaci- tance in order to reduce the voltage ripple or else disable burst mode operation by forcing continuous operation with the fcb pin. fault conditions: current limit and output shorts the ltc1625 current comparator can accommodate a maximum sense voltage of 150mv. this voltage and the sense resistance determine the maximum allowed peak inductor current. the corresponding output current limit is: i mv r i limit ds on t l = ()() 150 1 2 () r d the current limit value should be checked to ensure that i limit(min) > i o(max) . the minimum value of current limit generally occurs with the largest v in at the highest ambi- ent temperature, conditions which cause the highest power dissipation in the top mosfet. note that it is important to check for self-consistency between the assumed junction temperature of the top mosfet and the resulting value of i limit which heats the junction. caution should be used when setting the current limit based upon r ds(on) of the mosfets. the maximum current limit is determined by the minimum mosfet on- resistance. data sheets typically specify nominal and maximum values for r ds(on) , but not a minimum. a reasonable, but perhaps overly conservative, assumption is that the minimum r ds(on) lies the same amount below the typical value as the maximum r ds(on) lies above it. consult the mosfet manufacturer for further guidelines. the ltc1625 includes current foldback to help further limit load current when the output is shorted to ground. if the output falls by more than half, then the maximum sense voltage is progressively lowered from 150mv to 30mv. under short-circuit conditions with very low duty cycle, the ltc1625 will begin skipping cycles in order to limit the short-circuit current. in this situation the bottom mosfet r ds(on) will control the inductor current trough rather than the top mosfet controlling the inductor current peak. the short-circuit ripple current is deter- mined by the minimum on-time t on(min) of the ltc1625 (approximately 0.5 m s), the input voltage, and inductor value: d i l(sc) = t on(min) v in /l. the resulting short-circuit current is: i mv r i sc ds on bot t lsc = ()() + 30 1 2 ()( ) () r d normally, the top and bottom mosfets will be of the same type. a bottom mosfet with lower r ds(on) than the top may be chosen if the resulting increase in short-circuit current is tolerable. however, the bottom mosfet should never be chosen to have a higher nominal r ds(on) than the top mosfet. inductor core selection once the value for l is known, the type of inductor must be selected. high efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or kool m m ? cores. actual core loss is independent of core size for a fixed inductor value, but it is very dependent on the inductance selected. as inductance increases, core losses go down. unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. kool m m is a registered trademark of magnetics, inc.
12 ltc1625 applicatio n s i n for m atio n wu u u ferrite designs have very low core loss and are preferred at high switching frequencies, so design goals can con- centrate on copper loss and preventing saturation. ferrite core material saturates hard, which means that induc- tance collapses rapidly when the peak design current is exceeded. this results in an abrupt increase in inductor ripple current and consequent output voltage ripple. do not allow the core to saturate! molypermalloy (from magnetics, inc.) is a very good, low loss core material for toroids, but it is more expensive than ferrite. a reasonable compromise from the same manu- facturer is kool m m . toroids are very space efficient, especially when you can use several layers of wire. because they generally lack a bobbin, mounting is more difficult. however, designs for surface mount are available which do not increase the height significantly. schottky diode selection the schottky diode d1 shown in figure 1 conducts during the dead time between the conduction of the power mosfets. this prevents the body diode of the bottom mosfet from turning on and storing charge during the dead time, which could cost as much as 1% in efficiency. a 1a schottky diode is generally a good size for 3a to 5a regulators. the diode may be omitted if the efficiency loss can be tolerated. c in and c out selection in continuous mode, the drain current of the top mosfet is approximately a square wave of duty cycle v out /v in . to prevent large input voltage transients, a low esr input capacitor sized for the maximum rms current must be used. the maximum rms current is given by: ii v v v v rms o max out in in out @- ? ? ? ? () / 1 12 this formula has a maximum at v in = 2v out , where i rms = i o(max) /2. this simple worst-case condition is com- monly used for design because even significant deviations do not offer much relief. note that ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life. this makes it advisable to further derate the capacitor or to choose a capacitor rated at a higher temperature than required. several capacitors may also be placed in parallel to meet size or height requirements in the design. the selection of c out is primarily determined by the esr required to minimize voltage ripple. the output ripple d v out is approximately bounded by: dd v i esr fc out l out + ? ? ? ? 1 8 ()()( ) since d i l increases with input voltage, the output ripple is highest at maximum input voltage. typically, once the esr requirement is satisfied the capacitance is adequate for filtering and has the required rms current rating. manufacturers such as nichicon, united chemicon and sanyo should be considered for high performance through- hole capacitors. the os-con semiconductor dielectric capacitor available from sanyo has the lowest product of esr and size of any aluminum electrolytic at a somewhat higher price. in surface mount applications, multiple capacitors may have to be placed in parallel to meet the esr requirement. aluminum electrolytic and dry tantalum capacitors are both available in surface mount packages. in the case of tantalum, it is critical that the capacitors have been surge tested for use in switching power supplies. an excellent choice is the avx tps series of surface mount tantalum, available in case heights ranging from 2mm to 4mm. other capacitor types include sanyo os-con, nichicon pl se- ries, and sprague 593d and 595d series. consult the manufacturer for other specific recommendations. intv cc regulator an internal p-channel low dropout regulator produces the 5.2v supply which powers the drivers and internal cir- cuitry within the ltc1625. the intv cc pin can supply up to 50ma and must be bypassed to ground with a minimum of 4.7 m f tantalum or low esr electrolytic capacitance. good bypassing is necessary to supply the high transient currents required by the mosfet gate drivers.
13 ltc1625 applicatio n s i n for m atio n wu u u high input voltage applications in which large mosfets are being driven at high frequencies may cause the ltc1625 to exceed its maximum junction temperature rating. most of the supply current drives the mosfet gates unless an external extv cc source is used. the junction temperature can be estimated from the equations given in note 2 of the electrical characteristics. for example, the LTC1625Cgn is limited to less than 14ma from a 30v supply: t j = 70 c + (14ma)(30v)(130 c/w) = 125 c to prevent the maximum junction temperature from being exceeded, the input supply current must be checked when operating in continuous mode at high v in . extv cc connection the ltc1625 contains an internal p-channel mosfet switch connected between the extv cc and intv cc pins. whenever the extv cc pin is above 4.7v the internal 5.2v regulator shuts off, the switch closes and intv cc power is supplied via extv cc until extv cc drops below 4.5v. this allows the mosfet gate drive and control power to be derived from the output or other external source during normal operation. when the output is out of regulation (start-up, short circuit) power is supplied from the internal regulator. do not apply greater than 7v to the extv cc pin and ensure that extv cc v in . significant efficiency gains can be realized by powering intv cc from the output, since the v in current supplying the driver and control currents will be scaled by a factor of duty cycle/efficiency. for 5v regulators this simply means connecting the extv cc pin directly to v out . however, for 3.3v and other lower voltage regulators, additional cir- cuitry is required to derive intv cc power from the output. the following list summarizes the four possible connec- tions for extv cc : 1. extv cc left open (or grounded). this will cause intv cc to be powered from the internal 5.2v regulator resulting in an efficiency penalty of up to 10% at high input voltages. 2. extv cc connected directly to v out . this is the normal connection for a 5v regulator and provides the highest efficiency. 3. extv cc connected to an output-derived boost network. for 3.3v and other low voltage regulators, efficiency gains can still be realized by connecting extv cc to an output-derived voltage which has been boosted to greater than 4.7v. this can be done with either an inductive boost winding as shown in figure 5a or a capacitive charge pump as shown in figure 5b. 4. extv cc connected to an external supply. if an external supply is available in the 5v to 7v range (extv cc < v in ), it may be used to power extv cc providing it is compat- ible with the mosfet gate drive requirements. v in tk ltc1625 sgnd fcb extv cc tg sw optional extv cc connection 5v < v sec < 7v r3 r4 1625 f05a t1 1:n bg pgnd + c sec 1 m f v out v sec v in + c in 1n4148 + c out figure 5a: secondary output loop and extv cc connection v in tk ltc1625 extv cc v pump ? 2(v out ?v d ) tg sw 1625 f05b l1 bg pgnd + c out v out bat85 bat85 bat85 vn2222ll v in + c in + 1 m f 0.22 m f figure 5b: capacitive charge pump for extv cc
14 ltc1625 note that r ds(on) also varies with the gate drive level. if gate drives other than the 5.2v intv cc are used, this must be accounted for when selecting the mosfet r ds(on) . particular care should be taken with applications where extv cc is connected to the output. when the output voltage is between 4.7v and 5.2v, intv cc will be con- nected to the output and the gate drive is reduced. the resulting increase in r ds(on) will also lower the current limit. even applications with v out > 5.2v will traverse this region during start-up and must take into account the reduced current limit. topside mosfet driver supply (c b , d b ) an external bootstrap capacitor (c b in the functional diagram) connected to the boost pin supplies the gate drive voltage for the topside mosfet. this capacitor is charged through diode d b from intv cc when the sw node is low. note that the voltage across c b is about a diode drop below intv cc . when the top mosfet turns on, the switch node voltage rises to v in and the boost pin rises to approximately v in + intv cc . during dropout operation, c b supplies the top driver for as long as ten cycles between refreshes. thus, the boost capacitance needs to store about 100 times the gate charge required by the top mosfet. in many applications 0.22 m f is adequate. when adjusting the gate drive level , the final arbiter is the total input current for the regulator. if you make a change and the input current decreases, then you improved the efficiency. if there is no change in input current, then there is no change in efficiency. output voltage programming the ltc1625 has a pin selectable output voltage deter- mined by the v prog pin as follows: v prog = 0v v out = 3.3v v prog = intv cc v out = 5v v prog = open v out = adjustable remote sensing of the output voltage is provided by the v osense pin. for fixed 3.3v and 5v output applications an internal resistive divider is used and the v osense pin is connected directly to the output voltage as shown in figure 6a. when using an external resistive divider, the applicatio n s i n for m atio n wu u u v prog pin is left open and the v osense pin is connected to feedback resistors as shown in figure 6b. the output voltage is set by the divider as: vv r r out =+ ? ? ? ? 119 1 2 1 . v prog v out = 5v: intv cc v out = 3.3v: gnd ltc1625 v osense 1625 f06a sgnd c out v out + figure 6a. fixed 3.3v or 5v v out v prog open ltc1625 v osense 1625 f06b sgnd c out r1 r2 + figure 6b. adjustable v out run/soft start function the run/ss pin is a dual purpose pin that provides a soft start function and a means to shut down the ltc1625. soft start reduces surge currents from v in by gradually in- creasing the controllers current limit i th(max) . this pin can also be used for power supply sequencing. pulling the run/ss pin below 1.4v puts the ltc1625 into a low quiescent current shutdown (i q < 30 m a). this pin can be driven directly from logic as shown in figure 7. releas- ing the run/ss pin allows an internal 3 m a current source to charge up the external capacitor c ss . if run/ss has been pulled all the way to ground there is a delay before starting of approximately:
15 ltc1625 then v sec will droop. an external resistor divider from v sec to the fcb pin sets a minimum voltage v sec(min) : vv r r sec min () . @+ ? ? ? ? 119 1 4 3 if v sec drops below this level, the fcb voltage forces continuous operation until v sec is again above its minimum. minimum on-time considerations minimum on-time t on(min) is the smallest amount of time that the ltc1625 is capable of turning the top mosfet on and off again. it is determined by internal timing delays and the amount of gate charge required to turn on the top mosfet. low duty cycle applications may approach this minimum on-time limit and care should be taken to ensure that: t v vf on min out in () ()() < if the duty cycle falls below what can be accommodated by the minimum on-time, the ltc1625 will begin to skip cycles. the output voltage will continue to be regulated, but the ripple current and ripple voltage will increase. the minimum on-time for the ltc1625 is generally about 0.5 m s. however, as the peak sense voltage (i l(peak) ? r ds(on) ) decreases, the minimum on-time gradually increases up to about 0.7 m s. this is of particular concern in forced continuous applications with low ripple current at light loads. if the duty cycle drops below the minimum on-time limit in this situation, a significant amount of cycle skipping can occur with correspondingly larger current and voltage ripple. efficiency considerations the efficiency of a switching regulator is equal to the output power divided by the input power ( 100%). per- cent efficiency can be expressed as: %efficiency = 100% C (l1 + l2 + l3 + ...) applicatio n s i n for m atio n wu u u t v a csfc delay ss ss = m ? ? ? ? =m () 14 3 05 . ./ when the voltage on run/ss reaches 1.4v the ltc1625 begins operating with a clamp on i th at 0.8v. as the voltage on run/ss increases to approximately 3.1v, the clamp on i th is raised until its full 2.4v range is restored. this takes an additional 0.5s/ m f. during this time the load current will be folded back to approximately 30mv/r ds(on) until the output reaches half of its final value. diode d1 in figure 7 reduces the start delay while allowing c ss to charge up slowly for the soft start function. this diode and c ss can be deleted if soft start is not needed. the run/ss pin has an internal 6v zener clamp (see func- tional diagram). 3.3v or 5v run/ss d1 c ss 1625 f07 run/ss c ss figure 7. run/ss pin interfacing fcb pin operation when the fcb pin drops below its 1.19v threshold, continuous synchronous operation is forced. in this case, the top and bottom mosfets continue to be driven regardless of the load on the main output. burst mode operation is disabled and current reversal is allowed in the inductor. in addition to providing a logic input to force continuous operation, the fcb pin provides a means to regulate a flyback winding output. it can force continuous synchro- nous operation when needed by the flyback winding, regardless of the primary output load. the secondary output voltage v sec is normally set as shown in figure 5a by the turns ratio n of the transformer: v sec @ (n + 1)v out however, if the controller goes into burst mode operation and halts switching due to a light primary load current,
16 ltc1625 where l1, l2, etc. are the individual losses as a percentage of input power. it is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in ltc1625 circuits: 1. intv cc current. this is the sum of the mosfet driver and control currents. the driver current results from switching the gate capacitance of the power mosfets. each time a mosfet gate is switched on and then off, a packet of gate charge q g moves from intv cc to ground. the resulting current out of intv cc is typically much larger than the control circuit current. in continu- ous mode, i gatechg = f(q g(top) + q g(bot) ). by powering extv cc from an output-derived source, the additional v in current resulting from the driver and control currents will be scaled by a factor of duty cycle/ efficiency. for example, in a 20v to 5v application at 400ma load, 10ma of intv cc current results in ap- proximately 3ma of v in current. this reduces the loss from 10% (if the driver was powered directly from v in ) to about 3%. 2. dc i 2 r losses. since there is no separate sense resis- tor, dc i 2 r losses arise only from the resistances of the mosfets and inductor. in continuous mode the aver- age output current flows through l, but is chopped between the top mosfet and the bottom mosfet. if the two mosfets have approximately the same r ds(on) , then the resistance of one mosfet can simply be summed with the resistance of l to obtain the dc i 2 r loss. for example, if each r ds(on) = 0.05 w and r l = 0.15 w , then the total resistance is 0.2 w . this results in losses ranging from 2% to 8% as the output current increases from 0.5a to 2a for a 5v output. i 2 r losses cause the efficiency to drop at high output currents. 3. transition losses apply only to the topside mosfet, and only when operating at high input voltages (typi- cally 20v or greater). transition losses can be esti- mated from: transition loss = (1.7)(v in 2 )(i o(max) )(c rss )(f) applicatio n s i n for m atio n wu u u 4. ltc1625 v in supply current. the v in current is the dc supply current to the controller excluding mosfet gate drive current. total supply current is typically about 850 m a. if extv cc is connected to 5v, the ltc1625 will draw only 330 m a from v in and the remaining 520 m a will come from extv cc . v in current results in a small (< 1%) loss which increases with v in . other losses including c in and c out esr dissipative losses, schottky conduction losses during dead time and inductor core losses, generally account for less than 2% total additional loss. checking transient response the regulator loop response can be checked by looking at the load transient response. switching regulators take several cycles to respond to a step in dc (resistive) load current. when a load step occurs, v out immediately shifts by an amount equal to ( d i load )(esr), where esr is the effective series resistance of c out , and c out begins to charge or discharge. the regulator loop acts on the resulting feedback error signal to return v out to its steady- state value. during this recovery time v out can be moni- tored for overshoot or ringing which would indicate a stability problem. the i th pin external components shown in figure 1 will provide adequate compensation for most applications. a second, more severe transient is caused by connecting loads with large (> 1 m f) supply bypass capacitors. the discharged bypass capacitors are effectively put in parallel with c out , causing a rapid drop in v out . no regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. the only solution is to limit the rise time of the switch drive in order to limit the inrush current to the load. automotive considerations: plugging into the cigarette lighter as battery-powered devices go mobile, there is a natural interest in plugging into the cigarette lighter in order to conserve or even recharge battery packs during opera- tion. but before you connect, be advised: you are plug- ging into the supply from hell. the main battery line in an
17 ltc1625 applicatio n s i n for m atio n wu u u automobile is the source of a number of nasty potential transients, including load dump, reverse and double battery. load dump is the result of a loose battery cable. when the cable breaks connection, the field collapse in the alternator can cause a positive spike as high as 60v which takes several hundred milliseconds to decay. reverse battery is just what it says, while double battery is a consequence of tow truck operators finding that a 24v jump start cranks cold engines faster than 12v. the network shown in figure 8 is the most straightforward approach to protect a dc/dc converter from the ravages of an automotive battery line. the series diode prevents current from flowing during reverse battery, while the transient suppressor clamps the input voltage during load dump. note that the transient suppressor should not conduct during double-battery operation, but must still clamp the input voltage below breakdown of the converter. although the ltc1625 has a maximum input voltage of 36v, most applications will be limited to 30v by the mosfet v (br)dss . for 40% ripple current at maximum v in the inductor should be: l v khz a v v h 3 ? ? ? ? =m 33 225 0 4 2 1 33 22 16 . ( )( . )( ) . choosing a standard value of 15 m h results in a maximum ripple current of: d i v khz h v v a l max () . ()() . . = m ? ? ? ? = 33 225 15 1 33 22 083 next, check that the minimum value of the current limit is acceptable. assume a junction temperature close to a 70 c ambient with r 80 c = 1.3. i mv aa limit 3 w ? ? ? ? = 150 0 042 1 3 1 2 083 23 (. )(.) . . this is comfortably above i o(max) = 2a. now double-check the assumed t j : p v v a a pf khz mw mw mw top =w+ =+= 33 22 23 13 0042 1 7 22 2 3 180 225 43 77 120 2 2 . (. )(.)(. ) (.)( )(. )( )( ) t j = 70 c + (120mw)(50 c/w) = 76 c since r (76 c) @ r (80 c), the solution is self-consistent. a short circuit to ground will result in a folded back current of: i mv v s h a sc = w + ? ? ? ? m m = 30 003 11 1 2 15 0 5 15 12 (. )(.) ()(.) . with a typical value of r ds(on) and r (50 c) = 1.1. the resulting power dissipated in the bottom mosfet is: p vv v amw bot =w= 15 3 3 15 12 11 003 37 2 . (. )(.)(. ) which is less than under full load conditions. v in transient voltage suppressor general instrument 1.5ka24a 12v ltc1625 50a i pk rating 1625 f08 pgnd figure 8. automotive application protection design example as a design example, take a supply with the following specifications: v in = 12v to 22v (15v nominal), v out = 3.3v, i o(max) = 2a, and f = 225khz. the required r ds(on) can immediately be estimated: r mv a ds on () ()(.) . ==w 120 213 0 046 a 0.042 w siliconix si4412dy mosfet ( q ja = 50 c/w) is close to this value.
18 ltc1625 applicatio n s i n for m atio n wu u u + c ss 0.1 m f r c 10k c c1 470pf c c2 220pf m1 si4412dy c in 22 m f 35v 2 v in 12v to 22v v out 3.3v 2a m2 si4412dy d1 mbrs140t3 d b cmdsh-3 c vcc 4.7 m f 1625 f09 open intv cc v in tk extv cc ltc1625 sync v prog sw tg boost intv cc bg run/ss fcb i th sgnd v osense pgnd 1 2 3 4 5 6 7 8 16 15 14 13 12 11 10 9 c b 0.1 m f l1 15 m h + c out 100 m f 10v 0.065 w 2 + c in : avx tpse226m035r0300 c out : avx tpsd107m010r0065 l1: sumida cdrh125-150mc figure 9. 3.3v/2a fixed output at 225khz c in is chosen for an rms current rating of at least 1a at temperature. c out is chosen with an esr of 0.033 w for low output ripple. the output ripple in continuous mode will be highest at the maximum input voltage and is approximately: d v o = ( d i l(max) )(esr) = (0.83a)(0.033 w ) = 27mv the complete circuit is shown in figure 9. pc board layout checklist when laying out the printed circuit board, the following checklist should be used to ensure proper operation of the ltc1625. these items are also illustrated graphically in the layout diagram of figure 10. check the following in your layout: 1) connect the tk lead directly to the drain of the topside mosfet. then connect the drain to the (+) plate of c in . this capacitor provides the ac current to the top mosfet. 2) the power ground pin connects directly to the source of the bottom n-channel mosfet. then connect the source to the anode of the schottky diode and (C) plate of c in , which should have as short lead lengths as possible. 3) the ltc1625 signal ground pin must return to the (C) plate of c out . connect the (C) plate of c out to power ground at the source of the bottom mosfet 4) keep the switch node sw away from sensitive small- signal nodes. ideally the switch node should be placed on the opposite side of the power mosfets from the ltc1625. 5) connect the intv cc decoupling capacitor c vcc closely to the intv cc pin and the power ground pin. this capacitor carries the mosfet gate drive current. 6) does the v osense pin connect directly to the (+) plate of c out ? in adjustable applications, the resistive divider (r1, r2) must be connected between the (+) plate of c out and signal ground. place the divider near the ltc1625 in order to keep the high impedance v osense node short. 7) for applications with multiple switching power con- verters connected to the same v in , ensure that the input filter capacitance for the ltc1625 is not shared with the other converters. ac input current from another con- verter will cause substantial input voltage ripple that may interfere with proper operation of the ltc1625. a few inches of pc trace or wire ( ? 100nh) between c in and v in is sufficient to prevent sharing.
19 ltc1625 applicatio n s i n for m atio n wu u u + + c ss optional 5v extv cc connection m1 m2 d1 c vcc bold lines indicate high current paths 1625 f10 c in c out v in v out open open ext clk r1 output divider required with v prog open r2 c c1 r c c b d b + l1 + + v in tk extv cc ltc1625 sync v prog sw tg boost intv cc bg run/ss fcb i th sgnd v osense pgnd 2 1 3 4 5 6 7 8 16 15 14 13 12 11 10 9 figure 10. ltc1625 layout diagram typical applicatio n s u + c ss 0.1 m f r c 10k c c 330pf c in 15 m f 35v v in 5v to 28v v out 5v 1.2a m2 1/2 si9936dy m1 1/2 si9936dy c vcc 4.7 m f 1625 ta02 open intv cc c b 0.1 m f d b cmdsh-3 l1 39 m h c in : avx tpsd156m035r0300 c out : avx tpsd107m010r0100 l1: sumida cd104-390mc + c out 100 m f 10v 0.100 w + v in tk extv cc ltc1625 sync v prog sw tg boost intv cc bg run/ss fcb i th sgnd v osense pgnd 1 2 3 4 5 6 7 8 16 15 14 13 12 11 10 9 5v/1.2a fixed output at 225khz
20 ltc1625 typical applicatio n s u + c ss 0.1 m f r c 10k c c1 2.2nf c c2 220pf c in 10 m f 30v 3 c f 0.1 m f v in 5v to 28v v out 3.3v 7a m2 fds6680a d1 mbrs140t3 m1 fds6680a c vcc 4.7 m f 1625 ta05 open ext clk c in : sanyo 30sc10m c out : sanyo 6sa150m c b 0.22 m f d b cmdsh-3 l1 7 m h + c out 150 m f 6.3v 0.03 w 2 + r f 4.7 w v in tk extv cc ltc1625 sync v prog sw tg boost intv cc bg run/ss fcb i th sgnd v osense pgnd 1 2 3 4 5 6 7 8 16 15 14 13 12 11 10 9 2.5v/2.8a adjustable output + c ss 0.1 m f r c 10k c c1 1nf c c2 330pf c in 22 m f 35v 2 v in 5v to 28v v out 2.5v 2.8a m2 1/2 si4920dy d1 mbrs140t3 m1 1/2 si4920dy c vcc 4.7 m f 1625 ta03 open open c b 0.22 m f d b cmdsh-3 l1 15 m h r2 11k 1% c in : avx tpse226m020r0300 c out : avx tpsd107m010r0065 l1: sumida cdrh125-150mc + c f 0.1 m f r f 4.7 w c out 100 m f 10v 0.065 w 2 + r1 10k 1% v in tk extv cc ltc1625 sync v prog sw tg boost intv cc bg run/ss fcb i th sgnd v osense pgnd 1 2 3 4 5 6 7 8 16 15 14 13 12 11 10 9 3.3v/7a fixed output
21 ltc1625 typical applicatio n s u 3.3v/4a fixed output with 12v/120ma auxiliary output + c ss 0.1 m f r c 10k c c1 470pf c c2 220pf m2 irlr3103 c in : sanyo 30sc10m c out : avx tpsd107m010r0065 c sec : avx tajb335m035r t1: bh electronics 510-1079 *yes! use a standard recovery diode d1 mbrs140t3 m1 irlr3103 c vcc 4.7 m f 1625 ta04 ext clk c b 0.22 m f d b cmdsh-3 r f 4.7 w c f 0.1 m f c in 10 m f 30v 2 v in 6v to 20v c sec 3.3 m f 35v c1 0.01 m f m3 ndt410el d s sm4003tr* d2 cdmsh-3 t1 8 m h 1:2.53 r1 4.7k r4 95.3k 1% r3 11k 1% + + c out 100 m f 10v 0.065 w 3 v sec 12v 120ma v out 3.3v 4a + r s 100k c s 0.1 m f v in tk extv cc ltc1625 sync v prog sw tg boost intv cc bg run/ss fcb i th sgnd v osense pgnd 1 2 3 4 5 6 7 8 16 15 14 13 12 11 10 9 12v/2.2a adjustable output + c in : avx tpse226m020r0300 c out : avx tpse686m020r0150 l1: sumida cdrh127-270mc c ss 0.1 m f r c 22k c c 470pf c in 22 m f 35v 2 v in 12.5v to 28v v out 12v 2a r2 35.7k 1% r1 3.92k 1% m2 si4412dy m1 si4412dy c vcc 4.7 m f 1625ta06 open c b 0.1 m f d b cmdsh-3 l1 27 m h + c out 68 m f 20v 0.15 w 2 + v in tk extv cc ltc1625 sync v prog sw tg boost intv cc bg run/ss fcb i th sgnd v osense pgnd 1 2 3 4 5 6 7 8 16 15 14 13 12 11 10 9 c f 0.1 m f r f 4.7 w
22 ltc1625 typical applicatio n s u C 5v/4.5a positive to negative converter c in : sanyo 16sv220m c out : sanyo 6sv470m l1: magnetics kool-m 77120-a7, 9 turns, 17 gauge c ss 0.1 f r c 10k c c1 2.2nf v in 5v to 10v v out 5v 4.5a 1625ta08 d b cmdsh-3 m2 fds6670a l1 6 h c vcc 4.7 f c in 220 f 16v + + c out 470 f 6.3v + d 1 mbr140t3 v in tk extv cc ltc1625 sync v prog sw tg boost intv cc bg run/ss fcb i th sgnd v osense pgnd 1 2 3 4 5 6 7 8 16 15 14 13 12 11 10 9 c c2 220pf c f 0.1 f m1 fds6670a c b 0.22 f r f 4.7 c in : sanyo 20s68m c out : sanyo 16sa100m l1: 7a, 18 h kool-m 77120-a7, 15 turns, 17 gauge c ss 0.1 f r c 10k c c1 2.2nf v in 6v to 18v v out 12v 1625ta09 d b cmdsh-3 d4 bat85 l1 18 h c vcc 4.7 f c in 68 f 20v x2 + + c out 100 f 16v 30m x2 + d2 mbrs340t3 d5 bat85 d1 mbrs 340t3 z1 mmbz 5240 10v v in tk extv cc ltc1625 sync v prog sw tg boost intv cc bg run/ss fcb i th sgnd v osense pgnd 1 2 3 4 5 6 7 8 16 15 14 13 12 11 10 9 c c2 220pf c f 0.1 f m1 si4420dy m2 si4420dy m3 si4420dy m4 si4425dy c b 0.33 f r f 4.7 r1 3.92k r2 35.7k r1 100k 4 6 5 3 8 2 17 c1 470pf c2 0.1 f d3 bat85 1/2 ltc1693-2 1/2 ltc1693-2 v in 18 12 6 i out 4.0 3.3 2.0 single inductor, positive output buck boost
23 ltc1625 package descriptio n u dimensions in inches (millimeters) unless otherwise noted. 0.016 ?0.050 0.406 ?1.270 0.010 ?0.020 (0.254 ?0.508) 45 0 ?8 typ 0.008 ?0.010 (0.203 ?0.254) 1 2 3 4 5 6 7 8 0.150 ?0.157** (3.810 ?3.988) 16 15 14 13 0.386 ?0.394* (9.804 ?10.008) 0.228 ?0.244 (5.791 ?6.197) 12 11 10 9 s16 0695 0.053 ?0.069 (1.346 ?1.752) 0.014 ?0.019 (0.355 ?0.483) 0.004 ?0.010 (0.101 ?0.254) 0.050 (1.270) typ dimension does not include mold flash. mold flash shall not exceed 0.006" (0.152mm) per side dimension does not include interlead flash. interlead flash shall not exceed 0.010" (0.254mm) per side * ** s package 16-lead plastic small outline (narrow 0.150) (ltc dwg # 05-08-1610) gn package 16-lead plastic ssop (narrow 0.150) (ltc dwg # 05-08-1641) gn16 (ssop) 0398 * dimension does not include mold flash. mold flash shall not exceed 0.006" (0.152mm) per side ** dimension does not include interlead flash. interlead flash shall not exceed 0.010" (0.254mm) per side 12 3 4 5 6 7 8 0.229 ?0.244 (5.817 ?6.198) 0.150 ?0.157** (3.810 ?3.988) 16 15 14 13 0.189 ?0.196* (4.801 ?4.978) 12 11 10 9 0.016 ?0.050 (0.406 ?1.270) 0.015 0.004 (0.38 0.10) 45 0 ?8 typ 0.007 ?0.0098 (0.178 ?0.249) 0.053 ?0.068 (1.351 ?1.727) 0.008 ?0.012 (0.203 ?0.305) 0.004 ?0.0098 (0.102 ?0.249) 0.025 (0.635) bsc 0.009 (0.229) ref information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
24 ltc1625 ? linear technology corporation 1998 1625f lt/tp 1298 4k ? printed in usa part number description comments ltc1435a high efficiency synchronous step-down controller optimized for low duty cycle battery to cpu power applications ltc1436a-pll high efficiency low noise synchronous step-down controller pll synchronization and auxiliary linear regulator ltc1438 dual high efficiency step-down controller power-on reset and low-battery comparator ltc1530 high power synchronous step-down controller so-8 with current limit, no r sense saves space, fixed frequency ideal for 5v to 3.3v ltc1538-aux dual high efficiency step-down controller 5v standby output and auxiliary linear regulator ltc1649 3.3v input high power step-down controller 2.7v to 5v input, 90% efficiency, ideal for 3.3v to 1.xv C 2.xv up to 20a related parts linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 (408) 432-1900 l fax: (408) 434-0507 l www.linear-tech.com 3.3v/1.8a fixed output + c in : avx tpsd156m035r0300 c out : avx tpsd107m010r0100 l1: sumida cdrh125-270mc c ss 0.1 m f r c 10k c c1 1nf c c2 100pf c in 15 m f 35v 2 v in 5v to 28v v out 3.3v 1.8a m2 1/2 si4936dy d1 mbrs140t3 m1 1/2 si4936dy c vcc 4.7 m f 1625 ta07 open c b 0.1 m f d b cmdsh-3 l1 27 m h + c out 100 m f 10v 0.1 w 2 + v in tk extv cc ltc1625 sync v prog sw tg boost intv cc bg run/ss fcb i th sgnd v osense pgnd 1 2 3 4 5 6 7 8 16 15 14 13 12 11 10 9 typical applicatio n u

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